Crossover network

ABSTRACT

A peak limiter is described which employs the Hilbert transform. Effectively distinct processing occurs for low frequency and high frequency signals and harmonic distortion results only when the limiter is excited by high frequencies.

This is a divisional of application Ser. No. 480,921 filed Mar. 31,1983, now U.S. Pat. No. 4,495,643.

BACKGROUND OF THE INVENTION

1. Field of the Invention.

The invention relates to the field of audio peak limiters.

2. Prior Art.

Many techniques are used for peak limiting of signals including audiofrequency signals. Among them are fast-attack, slow-release automaticgain control (AGC) amplifiers, diode audio-frequency clippers, dioderadio-frequency clippers, and fast-attack slow-release AGC amplifiersemploying delay lines. Each technique has its own audible, subjectivestrengths and weaknesses.

Particularly when combined with delay line techniques, AGC techniquescan be configured to have no audible harmonic or intermodulationdistortion. However, they avoid such distortion by using relativelylittle non-linear modification of the audio waveform. Because such asystem only senses peak level (which has little or no correlation withloudness), if the peak-to-RMS ratio of the signal varies significantly,then highly unnatural loudness variations (sometimes called "pumping"and "hole-punching") can result.

Audio frequency clippers are extremely simple: a pair of back-to-backzener diodes which simply clip off any peaks exceeding a given thresholdis one example. This technique can severely modify the audio waveform ina non-linear way, producing audible harmonic and/or intermodulation (IM)distortion when used to excess. However, because this technique actsonly on instantaneous peaks, it causes no significant loudness variationand is therefore frequently used. Generally clipping is preceded byautomatic level control circuitry designed to control the amount ofdistortion produced by the clipping thereby limiting distortion toinaudible (or at least esthetically acceptable) levels. U.S. Pat. No.4,208,548 discloses a system for controlling the amount of distortionproduced by an audio frequency clipper.

The radio frequency (RF) clipper has proven popular for processing voicesignals in shortwave and communications applications. Here, the audiofrequency signal is modulated into a single-sideband suppressed-carrierRF signal. This RF signal is clipped and the clipped signal is thendemodulated. An interesting property of this technique is that noharmonic distortion is produced with a pure tone, since the firstharmonic produced by the clipping is located at an integral multiple ofthe RF carrier frequency. If the carrier frequency is 1 MHz, theharmonics occur at 2 MHz, 3 MHz, etc. These harmonics are eliminatedupon demodulation and ordinary filtering. (Upon demodulation, the 2 MHzharmonic becomes 1 MHz: still well outside the audio range.)

Unfortunately, with ordinary program material (consisting of manyfrequencies simultaneously), RF clipping can produce IM distortion whichis even more severe than that produced by audio frequency clipping. Thepeak level of the output must be instantaneously constrained to a givenlevel, thus causing the distortion. If no harmonic distortion ispermitted, the waveform modification necessary to control the peak levelmust be entirely at the expense of added IM distortion. Subjectively,this technique sounds far better than audio frequency clipping on voiceand substantially worse than audio frequency on music. This is becausewhen voice is clipped, the objectionable audible distortion is primarilyharmonic distortion in the frequency range above the fundamentalfrequencies of voice. These harmonics fall in the frequency range towhich the ear is most sensitive (1-5 kHz), and in which there is littlenaturally-occurring energy in voice to mask such harmonics. Accordingly,the harmonic-distortion suppression properties of the RF clipper arevery useful for voice.

In contrast, most music is much denser spectrally than is voice.Harmonic distortion can often add a pleasing brightness to music becausethe harmonics are harmoniously related to the music, and becausenaturally-occurring harmonics tend to mask the addition of moderateamounts of added harmonic distortion. But IM distortion is notharmonious, and therefore always degrades the subjective quality ofmusic. Therefore, audio frequency clipping (particularly very "hard"clipping characterized by a highly linear transfer curve up to theclipping threshold) tends to sound better than RF clipping on mostmusic. Exceptions occur with instruments having simple spectra with fewhigh-frequency components to mask unnaturally-induced harmonics.Examples of such instruments are grand piano, harp, nylon-stringedacoustic guitar, and fender-Rhodes electric piano.

The present invention behaves like an RF clipper below a certainfrequency (4 kHz in the preferred embodiment), and like an audiofrequency clipper above this frequency. Accordingly, no harmonicdistortion is produced by input material below 4 kHz and the advantagesof RF clipping are achieved on voice. "dull-sounding" instruments, andother such program material with little naturally-occurring highfrequency energy to mask harmonic distortion. Conversely, in the case ofmost music (particularly in a system employing high frequencypreemphasis, such that peak limiting must be effected after suchpreemphasis), most of the energy to be controlled by the peak limiter isusually located above 4 kHz. By processing this frequency band with theequivalent of audio frequency clipping, minimum difference-frequency IMdistortion (which is particularly objectionable when listened to afterhigh-frequency deemphasis in a receiver) is obtained.

The closest prior art known to Applicant is that shown in an articleentitled "Decomposition of Nonlinear Operators into `Harmonic`Components, with Applications to Audio Signal Processing", ElectronicsLetters, Vol. 12, No. 7, pp. 23-24 (Jan. 8, 1976) by M. A. Gerzon,(hereinafter referred to as "the Gerzon article"). With respect to thecrossover network of FIG. 6 of this application, the closest prior artknown to Applicant is "A Family of Linear-Phase Crossover Networks ofHigh Slope Derived by Time Delay", by Lipshitz and Vanderkooy, a paperpresented at the 69th Convention of the Audio Engineering Society (May12-15, 1981 at Los Angeles, Calif).

SUMMARY OF THE INVENTION

The limiter of the present invention performs an audio peak limitingfunction. The limiter effectively provides radio frequency clipping oflow frequencies and audio frequency clipping of high frequencies. Thus,little or no harmonic distortion occurs for voice whereas harmonicdistortion is permitted for high frequency signals. In the preferredembodiment the input signal is separated into two signals with 90degrees phase difference between the signals. That is, one signal is theHilbert transform of the other. The high frequency components areremoved from one of the signals. Both signals are coupled to a vectorsum generator, the output of which is "ORed" in a thresholding means.The output of the thresholding means provides a control signal forcontrolling the gain of a voltage-controlled amplifier. This amplifieris coupled to receive the audio signal. The vector sum generator withthe resultant control signal provides the functional equivalent of radiofrequency clipping for the low frequency components of the audio signal.The exclusion of the high frequency components from one of the inputs tothe vector sum generator prevents radio frequency clipping of the highfrequency components. The limiter for the high frequency componentsoperates as a feedforward circuit with infinite compression ratiolimiting using instantaneous attack and release times which isequivalent to an ordinary audio frequency clipper.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating one presently preferredembodiment of the limiter of the present invention.

FIG. 2 is a block diagram illustrating the preferred embodiment for thebandpass filter of FIG. 1.

FIG. 3a is a waveform used to explain the operation of the limiter ofFIG. 1.

FIG. 3b is a waveform used to explain the operation of the limiter ofFIG. 1.

FIG. 4 is a block diagram illustrating another preferred embodiment ofthe present invention where a second processing path is also employed.

FIG. 5 is a copy of FIG. 9 of U.S. Pat. No. 4,208,548. This figure isshown in this application since it is discussed in conjunction with thepresent invention.

FIG. 6 is a block diagram of a portion of the circuit of FIGS. 1 and 2which may be used as a crossover network.

DETAILED DESCRIPTION OF THE INVENTION

An audio peak limiter embodying a Hilbert transform generator isdescribed. In the following description, numerous specific details areset forth such as specific frequencies, etc. It will be obvious to oneskilled in the art, however, that the present invention may be practicedwithout these specific details. In other instances, well-known circuitshave been shown in block diagram form in order not to unnecessarilyobscure the present invention.

The present invention implements a Hilbert transform to help achieve theequivalent of radio frequency clipping. Such clipping is used for thelow frequency components of the audio signal without actual use of RFmodulation and demodulation. For a discussion of the implementation ofthis transform for a signal processing system, see the Gerzon article,particularly see FIG. 2(b) of that article. The Hilbert transform hasalso been employed to help generate single-sideband signals. See, forinstance, "Delay Lines Help Generate Quadrature Voice for SSB",Electronics, Apr. 13, 1978, beginning at p. 115 by Webb and Kelly.

Referring to FIG. 1, an input audio signal on line 10 is coupled tophase difference networks 5 and 6. These networks provide audio signalsat their outputs (lines 8 and 11, respectively) which are 90 degrees outof phase. The signal on line 11 is the Hilbert transform of the signalon line 8. The signal on line 11 is passed through a lowpass filter 14which is the presently preferred embodiment has cutoff frequency ofapproximately 4 kHz. (A 5th order Chebychev filter with a 0.1dB passbandripple is currently preferred.) A delay network 12 receives the outputfrom the network 5 (line 8) and provides a phase shift to match thephase shift of the filter 14. A phase match of approximately+or -1.5degrees is currently preferred, although performance is somewhatimproved by more accurate matching. The difference networks 5 and 6,delay network 12 and lowpass filter 14 are sometimes hereinafterreferred to as the "input means".

Note the signal on line 18 does not include the high frequencycomponents of the audio signal. This is quite distinct from theteachings of Gerzon where a single frequency band is employedthroughout.

In the input means the filter 14 may be placed before the differencenetwork 6, and also the delay network 12 may be placed before thedifference network 5. Moreover, the phase difference of 90 degreesbetween the outputs of networks 12 and 14 need be accurately maintainedonly in the frequency range where filter 14 produces significantresponse instead of throughout the entire frequency range processed bythe peak limiter of FIG. 1. (In a typical application, the input audiosignal on line 10 is first passed through a highpass filter with, forinstance, a 30 Hz cutoff.)

The vector sum generator 22 receives the signal on line 16 and line 18.The magnitude of these signals are squared, summed and the output online 26 represents the square root of this sum.

A thresholding circuit 29 for analog ORing is used. The circuit 29provides, as its output on line 32, the greater of the signal on line 26or a predetermined constant voltage threshold applied on line 28. Thesignal on line 32 provides a control signal for controlling the gain ofthe voltage-controlled amplifier (VCA) 30. (Also this signal, afterpassing through lowpass filter 38 provides a gain control signal for VCA34.) The VCA 30 receives audio signals (all frequency components) fromline 16.

The VCA 30 has a gain which is inversely proportional to the signal online 32. The square root function within the generator 22 can beeliminated if the gain of the VCA 30 (and VCA 34) has a response whichis inversely proportional to the square of the signal on line 32.

If the gain of VCA 30 is inversely proportional to the signal on line 32for the vector sum generator shown in FIG. 1, then VCA 30 provides thefunctional equivalent of radio frequency clipping for the low frequencycomponents of the audio signal. This is represented by the waveform inFIG. 3b, more specifically, an undistorted sinewave because no harmonicdistortion is produced and no IM distortion is possible for a singletone. The lack of distortion in FIG. 3b can be explained by recallingthe trigonometric identity √sin² ₇₄ +cos².sub.θ =1. This is what iscomputed by the vector sum generator 22. Since the output of thegenerator 22 is thus a constant, the control voltage on line 32 is alsoconstant and no distortion is generated by VCA 30.

It can be shown that if line 18 is disconnected from the vector sumgenerator 22, then the signal on line 40 is an audio frequency clippedsignal. In this case, the signals on lines 16 and 26 are identical and a"feedforward" infinite-compression-ratio limiter with instantaneousattack and release times is realized. This is equivalent to a simpleaudio frequency clipper. A sinewave for this audio frequency clipperwhich exceeds the threshold on line 28 is shown in FIG. 3a.

The signal on line 40 thus has the characteristics of an radio frequencyclipped signal for all signals below the cutoff frequency of filter 14,and has the characteristics of an audio frequency clipped signal for allsignals in the frequency range producing negligible outputs from filter14. In the transition region of filter 14, the output on line 18 ischaracterized by more and more harmonic distortion (and less and less IMdistortion, in the case of two or more frequencies) as the signal online 18 becomes smaller and smaller with increased frequency.

It is desirable that the peak levels on line 40 be the same for allfrequencies (both above and below the cutoff frequency of filter 14). Toassure this, the gain of filter 14 does not exceed the gain of network12, and the 90 degree phase relationship between the signals on line 16and 18 is maintained throughout all frequencies for which there is asignificant output from filter 14. For accurate RF clipping, the gainbetween the input signal on line 10 and the signal on line 16 should bethe same as the gain between the signal on line 10 and the signal online 18.

For the above-described system, filters 20 and 38, VCA 34, bandpassfilter 42, summer 48 and shelving filter 50 are not required.

An additional reduction in IM distortion can be achieved with additionalcomponents which more closely follow the teachings of the Gerzonarticle. A delay network identical to network 12 is coupled to receivethe signal on line 11. The output of this added network is connected toan additional VCA which is controlled by the voltage on line 32. As inthe Gerzon article, the output of this additional VCA is applied to oneinput terminal of another 90 degree phase difference network and theoutput of the VCA 30 is applied to the other input terminal of thisphase difference network. The two outputs of this phase differencenetwork are summed and this summation is the peak limited signal. Thisadds considerable complexity to the system and provides little advantageand moreover, requires the 90 degree phase difference network to have aconstant time delay to avoid distorting the peak levels. While this ispossible (see above-referenced Webb and Kelly article), it is quiteexpensive, particularly where a bucket brigade device or charge coupleddevice delay lines are employed. (A digital realization could also befabricated. However, with current costs of digital circuits, thisrealization would be even more expensive than an analog system.)

The above-described portions of FIG. 1 (without filters 20 and 38, VCA34, bandpass filter 40, summer 48, and shelving filter 50) can berefined employing the distortion-cancellation teachings of U.S. Pat. No.4,208,548 (hereinafter '548 patent). The peak limiting means 90 of FIG.5 (FIG. 9 of the '548 patent) can be replaced by the VCA 30 of FIG. 1;the input to peak limiting means 90 would be line 16. In this manner,the IM distortion-cancellation as described in the '548 patent would beobtained, although the output from amplifier 94 would have poorlycontrolled peak levels when compared with the control achieved by thepresent invention.

Consider a 100 Hz tone applied to the input (line 16) at a levelrequiring 10 dB of peak reduction. With the present invention, this gainreduction would occur without introducing harmonic distortion since 100Hz is well within the range of the radio frequency clipping. However,the output of the differential amplifier 92 of FIG. 5 must beconsidered. If the two inputs to this amplifier are a pair of 100 Hztones with 10 dB difference in level (due to the fact that the gain ofVCA 30 is -10 dB), the amplifier's output will be 6.7 dB above theoutput level of peak limiting means 90. When this is added back into thesignal, in the amplifier 94 of FIG. 5, a full 10 dB overshoot will occurand thus the effect of the gain reduction will be cancelled. Anadditional overshoot correction will now be required or safety clippingemployed, creating distortion. These problems are overcome in FIG. 1 ofthe present invention by including VCA 34, filters 20, 38, and 42, andsummer 48 which provide distortion cancellation.

Referring again to FIG. 1, the signal on line 16 is coupled to a secondVCA 34 through a lowpass filter 20 (line 24). VCA 30 is matched to VCA34 so that the gains will accurately track if the same control signal isapplied to both amplifiers. The output of the threshold circuit 29 iscoupled through a lowpass filter 38 which may be a 12 dB per octaveBessel filter with approximately 140 μsec. delay. The output of filter38, line 36, provides a control signal for VCA 34. Filter 38 removes thesharp "corners" in the control voltage making it less likely thataudible distortion will occur when modulating the audio signal.Unfortunately, this distortion reduction causes peaks to overshoot atthe output of the VCA 34. In particular, if a full bandwidth audiosignal is applied, there are many high frequency overshoots. Filter 20smooths the input to VCA 34, minimizing output overshoots which wouldotherwise occur from the smoothed control signal applied to VCA 34.

Referring back once again to FIG. 5, VCA 34 can be considered to be inseries with the "+" input of the amplfier 92. Recalling the 100 Hzexample discussed above, it can now be seen that because the controlvoltage is perfectly smooth in this case (no distortion is produced onthe sinewave at 100 Hz), filter 38 does not modify the control signalapplied to VCA 34. The control signal on lines 32 and 36 are identicaland the gains ofVCA 30 and 34 are identical as well. The signals at bothinput terminals of amplifier 92 are therefore identical, and no outputis produced at the output of amplifier 92. The level at the output ofamplifier 94 is therefore not disturbed, there are no overshoots, andthe undistorted RF clipping again is fully preserved.

With complex program material, the situation is more difficult tounderstand because the control signal is no longer constant. Filter 38introduces a 140 μsec. time delay into the control signal on line 36(some time delay is an inevitable side effect of lowpass filtering).Compensation for this time delay must be introduced into the audio pathto assure accurate distortion cancellation. Such additional time delaysare provided by filters 20 and 42. Filter 20 delays the input to VCA 34by 140 μsec., this corresponding to the delay associated with filter 38.Thus, both audio and control input signals to VCA 34 are delayed byidentical amounts of time. Since the output of VCA 30 precedes theoutput of VCA 34 by 140 μsec., it must be delayed 140 μsec. as well.This is done in a highpass filter or a bandpass filter 42. In this way,all the audio signals arrive at summing amplifier 48 at the same time,that is, with 140 μsec. delay.

It is interesting to note the manner in which distortion cancellation iseffected. Note first that the control voltage to VCA 34 is sufficientlysmooth to essentially eliminate audio distortion from VCA 34's output(line 44). The only audibly significant difference-frequency IMdistortion produced by the system is produced by VCA 30. Therefore, toeliminate all low frequency IM distortion produced by the limiter, theoutput of VCA 30 is highpassed or bandpassed. In the presently preferredembodiment, a bandpass filter 42 is employed, eliminating all lowfrequency components from VCA 30's output.

Filters 20 and 42 together form a constant-delay phase-linear crossovernetwork with steep slopes. Realization of such filters may be difficult,however, a realization is described in conjunction with FIG. 2. In FIG.2, filter 42 is shown as comprising filters 60 and 68, phase corrector72, lowpass filter 62 and a subtractor 76. The input to the network ofFIG. 2 is line 40; this being the input to the filter 42 of FIG. 1. Theoutput of the subtractor of FIG. 2, line 46 corresponds to the output ofthe filter 42 of FIG. 1.

In FIG. 2, the filters 60 and 68 and phase corrector 72 are all linear,and thus, may be placed in any order. The cascade of filter 60 and phasecorrector 72 may be either realized as a constant-delay lowpass filterwhere the output of the system is to be band limited, or an allpassnetwork approximating constant delay if system bandwidth limitation isnot required. In this case, filter 60 is eliminated, and the phasecorrector 72 becomes the constant-delay allpass network. However, abandwidth limitation is generally desired for broadcast or communicationapplications, making filter 42 of FIG. 1 a bandpass filter as opposed toa highpass filter.

The low frequency cutoff of the network of FIG. 2 (filter 42 of FIG. 1)is created by subtracting the output of the lowpass filter 62 from theoutput from the path comprising lowpass filter 60, shelving filter 68and the corrector 72 at the subtractor 76. The delay in this upper pathis selected to match the delay in the filter 62, which itself isdesigned to have approximately constant delay in its passband. Shelvingfilter 68 is fabricated to match the amplitude response in this upperpath to the high frequency rolloff characteristics of the filter 62.This operation is described in FIG. 10 of the '548 patent. Acomplementary shelving filter 50 is placed at the output of summer 48 ofFIG. 1 to restore a flat response to the output signal on line 52.

To understand the operation of the system, it should be noted thatfilters 20 and 62 of FIGS. 1 and 2 have identical responses. Assumefirst that the gain in the VCAs 30 and 34 are the same, then the signalson lines 44 and 66 (FIGS. 1 and 2, respectively) will also be identicalsince the signals pass through identical filters. However, thecontribution on line 66 is subtracted before the final output while thecontribution on line 44 is added. Therefore, in the case of identicalVCA gains, these signals will cancel completely, leaving the overallfrequency response of the system identical to the frequency responsethrough filters 60, 68, corrector 70 and the complementary shelvingfilter 50. This response will have either a lowpass or allpasscharacteristic depending upon whether filter 60 is used (i.e., dependingupon whether the system output is to be bandwidth-limited).

In typical operation, the gains of the VCAs 30 and 34 are almostidentical. The gains differ only for brief transients caused by theresponse of the lowpass filter 38, which as previously mentioned, isemployed to prevent introduction of audible IM distortion. Thus,distortion-free but gain-controlled low frequencies (containing someovershoots) pass to line 44 (output of VCA 34) while peak controlledhigh frequencies pass through filter 42. The summation on line 52contains essentially no IM distortion in the low frequency band butcontains some overshoots due to the overshoots inherent in thecombination of filters 60 and 72 of FIG. 2, and because of overshoots online 44. An overshoot corrector should be employed to correct for theseovershoots. This corrector can be a simple audio frequency clipper orwhere better bandwidth control is desired, can employ the overshootprotection circuit techniques described in U.S. Pat. No. 4,249,042.

Additional distortion reduction for the system of FIG. 1 can be achievedby operating in quadrature throughout the entire system and more fullyfollowing the teachings of the Gerzon article. This is shown in FIG. 4.

In FIG. 4 corresponding components and lines to those of FIG. 1 havebeen numbered with the same numerals. The added identical componentswithin FIG. 4 have been shown with a letter prefix. For instance, thedelay network 12a of FIG. 4 is identical to the delay network 12 ofFIGS. 1 and 4. This is also true for the lowpass filter 20a, VCAs 30aand 34a, bandpass filter 42a, and summer 48a. The output of the summers40 and 48a are coupled through the 90 degree phase difference networks60 and 61, and the output of these networks are summed in a summer 63.The output of the summer 63 is then coupled to a shelving filter 50;this filter having the complementary response to the shelving filter 68of FIG. 2, as previously mentioned.

The realization of Filter 42 as shown in FIG. 2 in combination with thefilter 20 of FIG. 1, as mentioned, form a constant-delay phase linearcrossover network with steep slopes. This network is useful for acrossover system for loudspeakers. (Note that the crossover frequency isdetermined by the lowpass filters 20 and 62 and thus, a crossoverfrequency can be selected which is different than the frequency bandscaused by the lowpass filter 14.) Such a crossover network is shown inFIG. 6.

In FIG. 6, the input audio signal on line 16 is shown coupled to a firstshelving filter 100. The output of the shelving filter is coupled to theinput of a complementary shelving filter 101 and a lowpass filter 102.The output of filter 101 is coupled to a delay network 103 with theoutput of this network being coupled to the plus terminal of asubtractor 104. The output of lowpass filter 102 is coupled to the minusterminal of the subtractor 104. The high frequency components arepresent at the output of the subtractor 104 and the low frequencycomponents at the output of the lowpass filter 102. The embodimentillustrated in FIG. 6 provides non-band limited signal. If the delaynetwork 103 is replaced with the cascade of a lowpass filter and phasecorrector (to provide linear phase), a band limited crossover network isrealized.

The network of FIG. 6 provides a flat summation of the lowpass andhighpass outputs as shown by the equations below; where A(s), B(s),D(s), H(s), and L(s) are transfer functions of s, a complex variable:##EQU1##

    L(s)=A(s) B(s)

Thus, H(s)+L(s)=D(s), a constant time delay.

With the embodiment of FIG. 6, an essentially linear-phase lowpassfilter 102 can be used while still achieving high slopes. Note that inthis configuration, shelving filter 101 in series with the delay network103, provides a non-flat response which is not normally perceived asbeing useful in a crossover network, but which is put to use in theembodiment of FIG. 6.

Thus, a peak limiting system has been disclosed which acts as a radiofrequency clipper for low frequencies, and an audio frequency clipperfor high frequencies. The output level of the system is preciselycontrolled for all frequencies including the transition frequenciesbetween the low frequency and high frequency bands which are selected.In the full system of FIG. 1, the audio signal is divided into twofrequency bands (not necessarily the same as the low frequency and highfrequency bands used for the radio frequency and audio frequencyclipping) for distortion cancellation. The difference-frequency IMdistortion produced is substantially eliminated, however, someovershoots beyond the desired peak output level occur.

While in the presently preferred embodiment, the entire system isrealized with analog circuits, portions or all of the above systems canbe realized with digital circuits. However, it is believed that with thecurrent cost and speed of digital circuits, a digital realization wouldbe more expensive than an analog realization.

I claim:
 1. A crossover network for an audio signal comprising:a firstshelving filter for receiving said audio signal; a second shelvingfilter coupled to the output of said first shelving filter; a lowpassfilter coupled to the output of said first shelving filter; a delaynetwork coupled to the output of said second shelving filter;subtracting means for subtracting two signals coupled to the output ofsaid delay network and said lowpass filter; whereby a crossover networkis realized with the high frequency band being present at the output ofsaid subtraction means and the low frequency band at the output of saidlowpass filter.
 2. The crossover network defined by claim 1 wherein saidfirst and second shelving filters have reciprocal characteristics.
 3. Aband limited crossover network for an audio signal comprising:a firstshelving filter receiving said audio signal; a second shelving filtercoupled to the output of said first sheliving filter; a first lowpassfilter coupled to the output of said first shelving filter; a secondlowpass filter and a phase corrector coupled to the output of saidsecond shelving filter; subtracting means for subtracting two signalscoupled to the output of said second lowpass filter and phase correctorand said first lowpass filter; whereby a band limited crossover networkis realized with a high frequency band being present at the output ofsaid subtraction means and low frequency band being present at theoutput of said first lowpass filter.
 4. The crossover network defined byclaim 3 wherein the first and second shelving filters have reciprocalcharacteristics.